High data rate CDMA wireless communication system

ABSTRACT

A novel and improved method and apparatus for high rate CDMA wireless communication is described. A set of individually gain adjusted subscriber channels are formed via the use of a set of orthogonal subchannel codes. The set of sub-channel codes are comprised of four Walsh codes. The pilot data and control data are combined onto one channel. The remaining two transmit channels are used for transmitting non-specified digital data including user data or signaling data, or both.

This application is a continuation of application Ser. No. 08/886,604, filed Jul. 1, 1997, now U.S. Pat. No. 6,396,804 issued May 28, 2002, entitled “High Data Rate CDMA Wireless Communication System which is a continuation in part of application Ser. No. 08/654,443, filed May 28, 1996, now U.S. Pat. No. 5,930,230, entitled “HIGH DATA RATE CDMA WIRELESS COMMUNICATION SYSTEM” issued Jul. 27, 1999.

BACKGROUND OF THE INVENTION

I. Field of the Invention

The present invention relates to communications. More particularly, the present invention relates to a novel and improved method and apparatus for high data rate CDMA wireless communication.

II. Description of the Related Art

Wireless communication systems including cellular, satellite and point to point communication systems use a wireless link comprised of a modulated radio frequency (RF) signal to transmit data between two systems. The use of a wireless link is desirable for a variety of reasons including increased mobility and reduced infrastructure requirements when compared to wire line communication systems. One drawback of using a wireless link is the limited amount of communication capacity that results from the limited amount of available RF bandwidth. This limited communication capacity is in contrast to wire based communication systems where additional capacity can be added by installing additional wire line connections.

Recognizing the limited nature of RF bandwidth, various signal processing techniques have been developed for increasing the efficiency with which wireless communication systems utilize the available RF bandwidth. One widely accepted example of such a bandwidth efficient signal processing technique is the IS-95 over the air interface standard and its derivatives such as IS-95-A (referred to hereafter collectively as the IS-95 standard) promulgated by the telecommunication industry association (TIA) and used primarily within cellular telecommunications systems. The IS-95 standard incorporates code division multiple access (CDMA) signal modulation techniques to conduct multiple communications simultaneously over the same RF bandwidth. When combined with comprehensive power control, conducting multiple communications over the same bandwidth increases the total number of calls and other communications that can be conducted in a wireless communication system by, among other things, increasing the frequency reuse in comparison to other wireless telecommunication technologies. The use of CDMA techniques in a multiple access communication system is disclosed in U.S. Pat. No. 4,901,307, entitled “SPREAD SPECTRUM COMMUNICATION SYSTEM USING SATELLITE OR TERRESTRIAL REPEATERS”, and U.S. Pat. No. 5,103,459, entitled “SYSTEM AND METHOD FOR GENERATING SIGNAL WAVEFORMS IN A CDMA CELLULAR TELEPHONE SYSTEM”, both of which are assigned to the assignee of the present invention and incorporated by reference herein.

FIG. 1 provides a highly simplified illustration of a cellular telephone system configured in accordance with the use of the IS-95 standard. During operation, a set of subscriber units 10 a-d conduct wireless communication by establishing one or more RF interfaces with one or more base stations 12 a-d using CDMA modulated RF signals. Each RF interface between a base station 12 and a subscriber unit 10 is comprised of a forward link signal transmitted from the base station 12, and a reverse link signal transmitted from the subscriber unit. Using these RF interfaces, a communication with another user is generally conducted by way of mobile telephone switching office (MTSO) 14 and public switch telephone network (PSTN) 16. The links between base stations 12, MTSO 14 and PSTN 16 are usually formed via wire line connections, although the use of additional RF or microwave links is also known.

In accordance with the IS-95 standard each subscriber unit 10 transmits user data via a single channel, non-coherent, reverse signal at a maximum data rate of 9.6 or 14.4 kbits/sec depending on which rate set from a set of rate sets is selected. A non-coherent link is one in which phase information is not utilized by the received system. A coherent link is one in which the receiver exploits knowledge of the carrier signals phase during processing. The phase information typically takes the form of a pilot signal, but can also be estimated from the data transmitted. The IS-95 standard calls for a set of sixty four Walsh codes, each comprised of sixty four chips, to be used for the forward link.

The use of a single channel, non-coherent, reverse link signal having a maximum data rate of 9.6 of 14.4 kbits/sec as specified by IS-95 is well suited for a wireless cellular telephone system in which the typical communication involves the transmission of digitized voice or lower rate digital data such a facsimile. A non-coherent reverse link was selected because, in a system in which up to 80 subscriber units 10 may communicate with a base station 12 for each 1.2288 MHz of bandwidth allocated, providing the necessary pilot data in the transmission from each subscriber unit 10 would substantially increase the degree to which a set of subscriber units 10 interfere with one another. Also, at data rates of 9.6 or 14.4 kbits/sec, the ratio of the transmit power of any pilot data to the user data would be significant, and therefore also increase inter-subscriber unit interference. The use of a single channel reverse link signal was chosen because engaging in only one type of communication at a time is consistent with the use of wireline telephones, the paradigm on which current wireless cellular communications is based. Also, the complexity of processing a single channel is less than that associated with processing multiple channels.

As digital communications progress, the demand for wireless transmission of data for applications such as interactive file browsing and video teleconferencing is anticipated to increase substantially. This increase will transform the way in which wireless communications systems are used, and the conditions under which the associated RF interfaces are conducted. In particular, data will be transmitted at higher maximum rates and with a greater variety of possible rates. Also, more reliable transmission may become necessary as errors in the transmission of data are less tolerable than errors in the transmission of audio information. Additionally, the increased number of data types will create a need to transmit multiple types of data simultaneously. For example, it may be necessary to exchange a data file while maintaining an audio or video interface. Also, as the rate of transmission from a subscriber unit increases the number of subscriber units 10 communicating with a base station 12 per amount of RF bandwidth will decrease, as the higher data transmission rates will cause the data processing capacity of the base station to be reached with fewer subscriber units 10. In some instances, the current IS-95 reverse link may not be ideally suited for all these changes. Therefore, the present invention is related to providing a higher data rate, bandwidth efficient, CDMA interface over which multiple types of communication can be performed.

SUMMARY OF THE INVENTION

A novel and improved method and apparatus for high rate CDMA wireless communication is described. In accordance with one embodiment of the invention, a set of individually gain adjusted subscriber channels are formed via the use of a set of orthogonal subchannel codes having a small number of PN spreading chips per orthogonal waveform period. Data to be transmitted via one of the transmit channels is low code rate error correction encoded and sequence repeated before being modulated with one of the subchannel codes, gain adjusted, and summed with data modulated using the other subchannel codes. The resulting summed data is modulated using a user long code and a pseudorandom spreading code (PN code) and upconverted for transmission. The use of the short orthogonal codes provides interference suppression while still allowing extensive error correction coding and repetition for time diversity to overcome the Raleigh fading commonly experienced in terrestrial wireless systems. In the exemplary embodiment of the invention provided, the set of sub-channel codes are comprised of four Walsh codes, each orthogonal to the remaining set and four chips in duration. In a preferred embodiment of the invention, two of the subscriber channel channels are multiplexed into a single traffic channel. The use of fewer traffic channels is preferred as it allows a smaller peak-to-average transmit power ratio. The use of different numbers of traffic channels is consistent with the invention.

In a first exemplary embodiment of the invention, pilot data is transmitted via a first one of the transmit channels and power control and other frame-by-frame control data are transmitted via a second transmit channel. In a preferred embodiment, the information on the pilot channel and the control subscriber channel, which includes the power control and frame-by-frame control data, are multiplex together onto one traffic channel to reduce the peak-to-average power ratio while still allowing for a continuous transmission. A continuous transmission is very desirable because it minimizes the possible interference with personal electronic equipment such as hearing aids and pacemakers. Since the pilot and control data are always transmitted, the resulting signal is still continuous. The other traffic channels are typically only active when the data of the type of that traffic channel is active. If the control data were multiplexed with a subscriber channel other than the pilot subscriber channel, the resulting traffic channel waveform would be discontinuous when the original traffic channel data is inactive. The other subscriber traffic channels could also be multiplexed into a single transmit channel. Two separate subscriber traffic channels are used here to allow for different gains and frame retransmission approaches for different types of traffic. The remaining two transmit channels are used for transmitting non-specified digital data including user data or signaling data, or both. In the exemplary embodiment, one of the two non-specified transmit channels is configured for BPSK modulation and the other for QPSK modulation. This is done to illustrate the versatility of the system. Both channels could be BPSK modulated or QPSK modulated in alternative embodiments of the invention.

Before modulation, the non-specified data is encoded where that encoding includes cyclic redundancy check (CRC) generation, convolutional encoding, interleaving, selective sequence repeating and BPSK or QPSK mapping. By varying the amount of repeating performed, and not restricting the amount of repeating to an integer number of symbol sequences, a wide variety of transmission rates including high data rates can be achieved. Furthermore, higher data rates can also be achieved by transmitting data simultaneously over both non-specified transmit channels. Also, by frequently updating the gain adjust performed on each transmit channel, the total transmit power used by the transmit system may be kept to a minimum such that the interference generated between multiple transmit systems is minimized, thereby increasing the overall system capacity.

BRIEF DESCRIPTION OF THE DRAWINGS

The features, objects, and advantages of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings in which like reference characters identify correspondingly throughout and wherein:

FIG. 1 is a block diagram of cellular telephone system;

FIG. 2 is a block diagram of a subscriber unit and base station configured in accordance with the exemplary embodiment of the invention;

FIG. 3 is a block diagram of a BPSK channel encoder and a QPSK channel encoder configured in accordance with the exemplary embodiment of the invention;

FIG. 4 is a block diagram of a transmit signal processing system configured in accordance with the exemplary embodiment of the invention;

FIG. 5 is a block diagram of a receive processing system configured in accordance with the exemplary embodiment of the invention;

FIG. 6 is a block diagram of a finger processing system configured in accordance with one embodiment of the invention;

FIG. 7 is a block diagram of a BPSK channel decoder and a QPSK channel decoder configured in accordance with the exemplary embodiment of the invention; and

FIG. 8 is a block diagram of the transmission system of the present invention wherein the control data and pilot data have been combined onto one channel;

FIG. 9 is a block diagram of the transmission system of the present invention wherein the control data and pilot data have been combined onto one channel including the filtering of the signals to be transmitted;

FIG. 10 is a receiver system of the present invention for receiving data wherein the power data and pilot data have been combined onto one channel.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

A novel and improved method and apparatus for high rate CDMA wireless communication is described in the context of the reverse link transmission portion of a cellular telecommunications system. While the invention is particularly adapted for use within the multipoint-to-point reverse link transmission of a cellular telephone system, the present invention is equally applicable to forward link transmissions. In addition, many other wireless communication systems will benefit by incorporation of the invention, including satellite based wireless communication systems, point to point wireless communication systems, and systems transmitting radio frequency signals via the use of co-axial or other broadband cables.

FIG. 2 is a block diagram of receive and transmit systems configured as a subscriber unit 100 and a base station 120 in accordance with one embodiment of the invention. A first set of data (BPSK data) is received by BPSK channel encoder 103, which generates a code symbol stream configured for performing BPSK modulation that is received by modulator 104. A second set of data (QPSK data) is received by QPSK channel encoder 102, which generates a code symbol stream configured for performing QPSK modulation that is also received by modulator 104. Modulator 104 also receives power control data and pilot data, which are modulated along with the BPSK and QPSK encoded data in accordance with code division multiple access (CDMA) techniques to generate a set of modulation symbols received by RF processing system 106. RF processing system 106 filters and upconverts the set of modulation symbols to a carrier frequency for transmission to the base station 120 using antenna 108. While only one subscriber unit 100 is shown, multiple subscriber units communicate with base station 120 in the preferred embodiment.

Within base station 120, RF processing system 122 receives the transmitted RF signals by way of antenna 121 and performs bandpass filtering, downconversion to baseband, and digitization. Demodulator 124 receives the digitized signals and performs demodulation in accordance with CDMA techniques to produce power control, BPSK, and QPSK soft decision data. BPSK channel decoder 128 decodes the BPSK soft decision data received from demodulator 124 to yield a best estimate of the BPSK data, and QPSK channel decoder 126 decodes the QPSK soft decision data received by demodulator 124 to produce a best estimate of the QPSK data. The best estimate of first and second set of data is then available for further processing or forwarding to a next destination, and the received power control data used either directly, or after decoding, to adjust the transmit power of the forward link channel used to transmit data to subscriber unit 100.

FIG. 3 is a block diagram of BPSK channel encoder 103 and QPSK channel encoder 102 when configured in accordance with the exemplary embodiment of the invention. Within BPSK channel encoder 103 the BPSK data is received by CRC check sum generator 130 which generates a check sum for each 20 ms frame of the first set of data. The frame of data along with the CRC check sum is received by tail bit generator 132 which appends tail bits comprised of eight logic zeros at the end of each frame to provide a known state at the end of the decoding process. The frame including the code tail bits and CRC check sum is then received by convolutional encoder 134 which performs, constraint length (K) 9, rate (R) ¼ convolutional encoding thereby generating code symbols at a rate four times the encoder input rate (E_(R)). In the alternative embodiment of the invention, other encoding rates are performed including rate ½, but the use of rate ¼ is preferred due to its optimal complexity-performance characteristics. Block interleaver 136 performs bit interleaving on the code symbols to provide time diversity for more reliable transmission in fast fading environments. The resulting interleaved symbols are received by variable starting point repeater 138, which repeats the interleaved symbol sequence a sufficient number of times N_(R) to provide a constant rate symbol stream, which corresponds to outputting frames having a constant number of symbols. Repeating the symbol sequence also increases the time diversity of the data to overcome fading. In the exemplary embodiment, the constant number of symbols is equal to 6,144 symbols for each frame making the symbol rate 307.2 kilosymbols per second (ksps). Also, repeater 138 uses a different starting point to begin the repetition for each symbol sequence. When the value of N_(R) necessary to generate 6,144 symbols per frame is not an integer, the final repetition is only performed for a portion of the symbol sequence. The resulting set of repeated symbols are received by BPSK mapper 139 which generates a BPSK code symbol stream (BPSK) of +1 and −1 values for performing BPSK modulation. In an alternative embodiment of the invention repeater 138 is placed before block interleaver 136 so that block interleaver 136 receives the same number of symbols for each frame.

Within QPSK channel encoder 102 the QPSK data is received by CRC check sum generator 140 which generates a check sum for each 20 ms frame. The frame including the CRC check sum is received by code tail bits generator 142 which appends a set of eight tail bits of logic zeros at the end of the frame. The frame, now including the code tail bits and CRC check sum, is received by convolutional encoder 144 which performs K=9, R=¼ convolutional encoding thereby generating symbols at a rate four times the encoder input rate (E_(R)). Block interleaver 146 performs bit interleaving on the symbols and the resulting interleaved symbols are received by variable starting point repeater 148. Variable starting point repeater 148 repeats the interleaved symbol sequence a sufficient number of times N_(R) using a different starting point within the symbol sequence for each repetition to generate 12,288 symbols for each frame making the code symbol rate 614.4 kilosymbols per second (ksps). When N_(R) is not an integer, the final repetition is performed for only a portion of the symbol sequence. The resulting repeated symbols are received by QPSK mapper 149 which generates a QPSK code symbol stream configured for performing QPSK modulation comprised of an in-phase QPSK code symbol stream of +1 and −1 values (QPSK_(I)), and a quadrature-phase QPSK code symbol stream of +1 and −1 values (QPSK_(Q)). In an alternative embodiment of the invention repeater 148 is placed before block interleaver 146 so that block interleaver 146 receives the same number of symbols for each frame.

FIG. 4 is a block diagram of modulator 104 of FIG. 2 configured in accordance with the exemplary embodiment of the invention. The BPSK symbols from BPSK channel encoder 103 are each modulated by Walsh code W₂ using a multiplier 150 b, and the QPSK_(I) and QPSK_(Q) symbols from QPSK channel encoder 102 are each modulated with Walsh code W₃ using multipliers 150 c and 150 d. The power control data (PC) is modulated by Walsh code W₁ using multiplier 150 a. Gain adjust 152 receives pilot data (PILOT), which in the preferred embodiment of the invention is comprised of the logic level associated with positive voltage, and adjusts the amplitude according to a gain adjust factor A₀. The PILOT signal provides no user data but rather provides phase and amplitude information to the base station so that it can coherently demodulate the data carried on the remaining sub-channels, and scale the soft-decision output values for combining. Gain adjust 154 adjusts the amplitude of the Walsh code W₁, modulated power control data according to gain adjust factor A₁, and gain adjust 156 adjusts the amplitude of the Walsh code W₂ modulated BPSK channel data according amplification variable A₂. Gain adjusts 158 a and b adjust the amplitude of the in-phase and quadrature-phase Walsh code W₃ modulated QPSK symbols respectively according to gain adjust factor A₃. The four Walsh codes used in the preferred embodiment of the invention are shown in Table I.

TABLE I Modulation Walsh Code Symbols W₀ + + + + W₁ + − + − W₂ + + − − W₃ + − − +

It will be apparent to one skilled in the art that the W₀ code is effectively no modulation at all, which is consistent with processing of the pilot data shown. The power control data is modulated with the W₁ code, the BPSK data with the W₂ code, and the QPSK data with the W₃ code. Once modulated with the appropriate Walsh code, the pilot, power control data, and BPSK data are transmitted in accordance with BPSK techniques, and QPSK data (QPSK_(I) and QPSK_(Q)) in accordance with QPSK techniques as described below. It should also be understood that it is not necessary that every orthogonal channel be used, and that the use of only three of the four Walsh codes where only one user channel is provided is employed in an alternative embodiment of the invention.

The use of short orthogonal codes generates fewer chips per symbol, and therefore allows for more extensive coding and repetition when compared to systems incorporating the use of longer Walsh codes. This more extensive coding and repetition provides protection against Raleigh fading which is a major source of error in terrestrial communication systems. The use of other numbers of codes and code lengths is consistent with the present invention, however, the use of a larger set of longer Walsh codes reduces this enhanced protection against fading. The use of four chip codes is considered optimal because four channels provides substantial flexibility for the transmission of various types of data as illustrated below while also maintaining short code length.

Summer 160 sums the resulting amplitude adjusted modulation symbols from gain adjusts 152, 154, 156 and 158 a to generate summed modulation symbols 161. PN spreading codes PN_(I) and PN_(Q) are spread via multiplication with long code 180 using multipliers 162 a and b. The resulting pseudorandom code provided by multipliers 162 a and 162 b are used to modulate the summed modulation symbols 161, and gain adjusted quadrature-phase symbols QPSK_(Q) 163, via complex multiplication using multipliers 164 a-d and summers 166 a and b. The resulting in-phase term X_(I) and quadrature-phase term X_(Q) are then filtered (filtering not shown), and upconverted to the carrier frequency within RF processing system 106 shown in a highly simplified form using multipliers 168 and an in-phase and a quadrature-phase sinusoid. An offset QPSK upconversion could also be used in an alternative embodiment of the invention. The resulting in-phase and quadrature-phase upconverted signals are summed using summer 170 and amplified by master amplifier 172 according to master gain adjust A_(M) to generate signal s(t) which is transmitted to base station 120. In the preferred embodiment of the invention, the signal is spread and filtered to a 1.2288 MHz bandwidth to remain compatible with the bandwidth of existing CDMA channels.

By providing multiple orthogonal channels over which data may be transmitted, as well as by using variable rate repeaters that reduce the amount of repeating N_(R) performed in response to high input data rates, the above described method and system of transmit signal processing allows a single subscriber unit or other transmit system to transmit data at a variety of data rates. In particular, by decreasing the rate of repetition N_(R) performed by variable starting point repeaters 138 or 148 of FIG. 3, an increasingly higher encoder input rate E_(R) can be sustained. In an alternative embodiment of the invention rate ½ convolution encoding is performed with the rate of repetition N_(R) increased by two. A set of exemplary encoder rates E_(R) supported by various rates of repetition N_(R) and encoding rates R equal to ¼ and ½ for the BPSK channel and the QPSK channel are shown in Tables II and III respectively.

TABLE II BPSK Channel N_(R,R=1/4) N_(R,R=1/2) (Repe- (Repe- Encoder Out tition Encoder Out tition E_(R,BPSK) R = 1/4 Rate, R = 1/2 Rate, Label (bps) (bits/frame) R = 1/4) (bits/frame) R = 1/2) High 76,800 6,144  1 3,072  2 Rate-72 High 70,400 5,632  1 1/11 2,816  2 2/11 Rate-64 51,200 4,096  1 1/2 2,048  3 High 38,400 3,072  2 1,536  4 Rate-32 25,600 2,048  3 1,024  6 RS2-Full 14,400 1,152  5 1/3   576  10 2/3 Rate RS1-Full  9,600   768  8   384  16 Rate NULL   850   68 90 6/17   34 180 12/17

TABLE III QPSK Channel N_(R,R=1/4) N_(R,R=1/2) (Repe- (Repe- Encoder Out tition Encoder Out tition E_(R,QPSK) R = 1/4 Rate, R = 1/2 Rate, Label (bps) (bits/frame) R = 1/4) (bits/frame) R = 1/2) 153,600 12,288   1 6,144  2 High 76,800 6,144  2 3,072  4 Rate-72 High 70,400 5,632  2 2/11 2,816  4 4/11 Rate-64 51,200 4,096  3 2,048  6 High 38,400 3,072  4 1,536  8 Rate-32 25,600 2,048  6 1,024  12 RS2-Full 14,400 1,152  10 2/3   576  21 1/3 Rate RS1-Full  9,600   768  16   384  32 Rate NULL   850   68 180 12/17   34 361 7/17

Tables II and III show that by adjusting the number of sequence repetitions N_(R), a wide variety of data rates can be supported including high data rates, as the encoder input rate E_(R) corresponds to the data transmission rate minus a constant necessary for the transmission of CRC, code tail bits and any other overhead information. As also shown by tables II and III, QPSK modulation may also be used to increase the data transmission rate. Rates expected to be used commonly are provided labels such as “High Rate-72” and “High Rate-32.” Those rates noted as High Rate-72, High Rate-64, and High Rate-32 have traffic rates of 72, 64 and 32 kbps respectively, plus multiplexed in signaling and other control data with rates of 3.6, 5.2, and 5.2 kbps respectively, in the exemplary embodiment of the invention. Rates RS1-Full Rate and RS2-Full Rate correspond to rates used in IS-95 compliant communication systems, and therefore are also expected to receive substantial use for purposes of compatibility. The null rate is the transmission of a single bit and is used to indicate a frame erasure, which is also part of the IS-95 standard.

The data transmission rate may also be increased by simultaneously transmitting data over two or more of the multiple orthogonal channels performed either in addition to, or instead of, increasing the transmission rate via reduction of the repetition rate N_(R). For example, a multiplexer (not shown) could split a single data source into a multiple data sources to be transmitted over multiple data sub-channels. Thus, the total transmit rate can be increased via either transmission over a particular channel at higher rates, or multiple transmission performed simultaneously over multiple channels, or both, until the signal processing capability of the receive system is exceeded and the error rate becomes unacceptable, or the maximum transmit power of the transmit system power is reached.

Providing multiple channels also enhances flexibility in the transmission of different types of data. For example, the BPSK channel may be designated for voice information and the QPSK channel designated for transmission of digital data. This embodiment could be more generalized by designating one channel for transmission of time sensitive data such as voice at a lower data rate, and designating the other channel for transmission of less time sensitive data such as digital files. In this embodiment interleaving could be performed in larger blocks for the less time sensitive data to further increase time diversity. In another embodiment of the invention, the BPSK channel performs the primary transmission of data, and the QPSK channel performs overflow transmission. The use of orthogonal Walsh codes eliminates or substantially reduces any interference among the set of channels transmitted from a subscriber unit, and thus minimizes the transmit energy necessary for their successful reception at the base station.

To increase the processing capability at the receive system, and therefore increase the extent to which the higher transmission capability of the subscriber unit may be utilized, pilot data is also transmitted via one of the orthogonal channels. Using the pilot data, coherent processing can be performed at the receive system by determining and removing the phase offset of the reverse link signal. Also, the pilot data can be used to optimally weigh multipath signals received with different time delays before being combined in a rake receiver. Once the phase offset is removed, and the multipath signals properly weighted, the multipath signals can be combined decreasing the power at which the reverse link signal must be received for proper processing. This decrease in the required receive power allows greater transmission rates to be processed successfully, or conversely, the interference between a set of reverse link signals to be decreased. While some additional transmit power is necessary for the transmission of the pilot signal, in the context of higher transmission rates the ratio of pilot channel power to the total reverse link signal power is substantially lower than that associated with lower data rate digital voice data transmission cellular systems. Thus, within a high data rate CDMA system the E_(b)/N₀ gains achieved by the use of a coherent reverse link outweigh the additional power necessary to transmit pilot data from each subscriber unit.

The use of gain adjusts 152-158 as well as master amplifier 172 further increases the degree to which the high transmission capability of the above described system can be utilized by allowing the transmit system to adapt to various radio channel conditions, transmission rates, and data types. In particular, the transmit power of a channel that is necessary for proper reception may change over time, and with changing conditions, in a manner that is independent of the other orthogonal channels. For example, during the initial acquisition of the reverse link signal the power of the pilot channel may need to be increased to facilitate detection and synchronization at the base station. Once the reverse link signal is acquired, however, the necessary transmit power of the pilot channel would substantially decrease, and would vary depending on various factors including the subscriber units rate of movement. Accordingly, the value of the gain adjust factor A₀ would be increased during signal acquisition, and then reduced during an ongoing communication. In another example, when information more tolerable of error is being transmitted via the forward link, or the environment in which the forward link transmission is taking place is not prone to fade conditions, the gain adjust factor A₁ may be reduced as the need to transmit power control data with a low error rate decreases. In one embodiment of the invention, whenever power control adjustment is not necessary the gain adjust factor A₁ is reduced to zero.

In another embodiment of the invention, the ability to gain adjust each orthogonal channel or the entire reverse link signal is further exploited by allowing the base station 120 or other receive system to alter the gain adjust of a channel, or of the entire reverse link signal, via the use of power control commands transmitted via the forward link signal. In particular, the base station may transmit power control information requesting the transmit power of a particular channel or the entire reverse link signal be adjusted. This is advantageous in many instances including when two types of data having different sensitivity to error, such as digitized voice and digital data, are being transmitted via the BPSK and QPSK channels. In this case, the base station 120 would establish different target error rates for the two associated channels. If the actual error rate of a channel exceeded the target error rate, the base station would instruct the subscriber unit to reduce the gain adjust of that channel until the actual error rate reached the target error rate. This would eventually lead to the gain adjust factor of one channel being increased relative to the other. That is, the gain adjust factor associated with the more error sensitive data would be increased relative to the gain adjust factor associated with the less sensitive data. In other instances, the transmit power of the entire reverse link may require adjustment due to fade conditions or movement of the subscriber unit 100. In these instances, the base station 120 can do so via transmission of a single power control command.

Thus, by allowing the gain of the four orthogonal channels to be adjusted independently, as well as in conjunction with one another, the total transmit power of the reverse link signal can be kept at the minimum necessary for successful transmission of each data type, whether it is pilot data, power control data, signaling data, or different types of user data. Furthermore, successful transmission can be defined differently for each data type. Transmitting with the minimum amount of power necessary allows the greatest amount of data to be transmitted to the base station given the finite transmit power capability of a subscriber unit, and also reduces the interference between subscriber units. This reduction in interference increases the total communication capacity of the entire CDMA wireless cellular system.

The power control channel used in the reverse link signal allows the subscriber unit to transmit power control information to the base station at a variety of rates including a rate of 800 power control bits per second. In the preferred embodiment of the invention, a power control bit instructs the base station to increase or decrease the transmit power of the forward link traffic channel being used to transmit information to the subscriber unit. While it is generally useful to have rapid power control within a CDMA system, it is especially useful in the context of higher data rate communications involving data transmission, because digital data is more sensitive to errors, and the high transmission causes substantial amounts of data to be lost during even brief fade conditions. Given that a high speed reverse link transmission is likely to be accompanied by a high speed forward link transmission, providing for the rapid transmission of power control over the reverse link further facilitates high speed communications within CDMA wireless telecommunications systems.

In an alternative exemplary embodiment of the invention a set of encoder input rates E_(R) defined by the particular N_(R) are used to transmit a particular type of data. That is, data may be transmitted at a maximum encoder input rate E_(R) or at a set of lower encoder input rates E_(R), with the associated N_(R) adjusted accordingly. In the preferred implementation of this embodiment, the maximum rates corresponds to the maximum rates used in IS-95 compliant wireless communication system, referred to above with respect to Tables II and III as RS1-Full Rate and RS2-Full Rate, and each lower rate is approximately one half the next higher rate, creating a set of rates comprised of a full rate, a half rate, a quarter rate, and an eighth rate. The lower data rates are preferable generated by increasing the symbol repetition rate N_(R) with value of N_(R) for rate set one and rate set two in a BPSK channel provided in Table IV.

TABLE IV RS1 and RS2 Rate Sets in BPSK Channel N_(R,R=1/4) N_(R,R=1/2) (Repe- (Repe- Encoder Out tition Encoder Out tition E_(R,QPSK) R = 1/4 Rate, R = 1/2 Rate, Label (bps) (bits/frame) R = 1/4) (bits/frame) R = 1/2) RS2-Full 14,400  1,152    5 1/3 576  10 2/3 Rate RS2-Half 7,200 576 10 2/3 288  21 1/3 Rate RS2-Quar- 3,600 288 21 1/3 144  42 2/3 ter Rate RS2- 1,900 152 40 8/19  76  80 16/19 Eighth Rate RS1-Full 9,600 768  8 384  16 Rate RS1-Half 4,800 384 16 192  32 Rate RS1-Quar- 2,800 224 27 3/7 112  54 6/7 ter Rate RS1- 1,600 128 48  64  96 Eighth Rate NULL   850  68 90 6/17  34 180 12/17

The repetition rates for a QPSK channel is twice that for the BPSK channel.

In accordance with the exemplary embodiment of the invention, when the data rate of a frame changes with respect to the previous frame the transmit power of the frame is adjusted according to the change in transmission rate. That is, when a lower rate frame is transmitted after a higher rate frame, the transmit power of the transmit channel over which the frame is being transmitted is reduced for the lower rate frame in proportion to the reduction in rate, and vice versa. For example, if the transmit power of a channel during the transmission of a full rate frame is transmit power T, the transmit power during the subsequent transmission of a half rate frame is transmit power T/2. The reduction in transmit power is preferably performed by reducing the transmit power for the entire duration of the frame, but may also be performed by reducing the transmit duty cycle such that some redundant information is “blanked out.” In either case, the transmit power adjustment takes place in combination with a closed loop power control mechanism whereby the transmit power is further adjusted in response to power control data transmitted from the base station.

FIG. 5 is a block diagram of RF processing system 122 and demodulator 124 of FIG. 2 configured in accordance with the exemplary embodiment of the invention. Multipliers 180 a and 180 b dowconvert the signals received from antenna 120 with an in-phase sinusoid and a quadrature phase sinusoid producing in-phase receive samples R_(I) and quadrature-phase receive samples R_(Q) receptively. It should be understood that RF processing system 122 is shown in a highly simplified form, and that the signals are also match filtered and digitized (not shown) in accordance with widely known techniques. Receive samples R_(I) and R_(Q) are then applied to finger demodulators 182 within demodulator 124. Each finger demodulator 182 processes an instance of the reverse link signal transmitted by subscriber unit 100, if such an instance is available, where each instance of the reverse link signal is generated via multipath phenomenon. While three finger demodulators are shown, the use of alternative numbers of finger processors are consistent with the invention including the use of a single finger demodulator 182. Each finger demodulator 182 produces a set of soft decision data comprised of power control data, BPSK data, and QPSK_(I), data and QPSK_(Q) data. Each set of soft decision data is also time adjusted within the corresponding finger demodulator 182, although time adjustment could be performed within combiner 184 in an alternative embodiment of the invention. Combiner 184 then sums the sets of soft decision data received from finger demodulators 182 to yield a single instance of power control, BPSK, QPSK_(I) and QPSK_(Q) soft decision data.

FIG. 6 is block diagram a finger demodulator 182 of FIG. 5 configured in accordance with the exemplary embodiment of the invention. The R_(I) and R_(Q) receive samples are first time adjusted using time adjust 190 in accordance with the amount of delay introduced by the transmission path of the particular instance of the reverse link signal being processed. Long code 200 is mixed with pseudorandom spreading codes PN_(I) and PN_(Q) using multipliers 201, and the complex conjugate of the resulting long code modulated PN_(I) and PN_(Q) spreading codes are complex multiplied with the time adjusted R_(I) and R_(Q) receive samples using multipliers 202 and summers 204 yielding terms X_(I) and X_(Q). Three separate instances of the X_(I) and X_(Q) terms are then demodulated using the Walsh codes W₁, W₂ and W₃ respectively, and the resulting Walsh demodulated data is summed over four demodulation chips using 4 to 1 summers 212. A fourth instance of the X_(I) and X_(Q) data is summed over four demodulation chips using summers 208, and then filtered using pilot filters 214. In the preferred embodiment of the invention pilot filter 214 performs averaging over a series of summations performed by summers 208, but other filtering techniques will be apparent to one skilled in the art. The filtered in-phase and quadrature-phase pilot signals are used to phase rotate and scale the W₁, and W₂ Walsh code demodulated data in accordance with BPSK modulated data via complex conjugate multiplication using multipliers 216 and adders 217 yielding soft decision power control and BPSK data. The W₃ Walsh code modulated data is phase rotated using the in-phase and quadrature-phase filtered pilot signals in accordance with QPSK modulated data using multipliers 218 and adders 220, yielding soft decision QPSK data. The soft decision power control data is summed over 384 modulation symbols by 384 to 1 summer 222 yielding power control soft decision data. The phase rotated W₂ Walsh code modulated data, the W₃ Walsh code modulated data, and the power control soft decision data are then made available for combining. In an alternative embodiment of the invention, encoding and decoding is performed on the power control data as well.

In addition to providing phase information the pilot may also be used within the receive system to facilitate time tracking. Time tracking is performed by also processing the received data at one sample time before (early), and one sample time after (late), the present receive sample being processed. To determine the time that most closely matches the actual arrival time, the amplitude of the pilot channel at the early and late sample time can be compared with the amplitude at the present sample time to determine that which is greatest. If the signal at one of the adjacent sample times is greater than that at the present sample time, the timing can be adjusted so that the best demodulation results are obtained.

FIG. 7 is a block diagram of BPSK channel decoder 128 and QPSK channel decoder 126 (FIG. 2) configured in accordance with the exemplary embodiment of the invention. BPSK soft decision data from combiner 184 (FIG. 5) is received by accumulator 240 which stores the first sequence of 6,144/N_(R) demodulation symbols in the received frame where N_(R) depends on the transmission rate of the BPSK soft decision data as described above, and adds each subsequent set of 6,144/N_(R) demodulated symbols contained in the frame with the corresponding stored accumulated symbols. Block deinterleaver 242 deinterleaves the accumulated soft decision data from accumulator 240, and Viterbi decoder 244 decodes the deinterleaved soft decision data to produce hard decision data as well as CRC check sum results. Within QPSK decoder 126 QPSK_(I) and QPSK_(Q) soft decision data from combiner 184 (FIG. 5) are demultiplexed into a single soft decision data stream by demux 246 and the single soft decision data stream is received by accumulator 248 which accumulates every 6,144/N_(R) demodulation symbols where N_(R) depends on the transmission rate of the QPSK data. Block deinterleaver 250 deinterleaves the soft decision data from accumulator 248, and Viterbi decoder 252 decodes the deinterleaved modulation symbols to produce hard decision data as well as CRC check sum results. In the alternative exemplary embodiment described above with respect to FIG. 3 in which symbol repetition is performed before interleaving, accumulators 240 and 248 are placed after block deinterleavers 242 and 250. In the embodiment of the invention incorporating the use of rate sets, and therefore in which the rate of particular frame is not known, multiple decoders are employed, each operating at a different transmission rate, and then the frame associated with the transmission rate most likely to have been used is selected based on the CRC checksum results. The use of other error checking methods is consistent with the practice of the present invention.

Now turning to FIG. 8, a reverse link transmission system in which the control data and the pilot data have been combined onto one channel is illustrated. It should be noted that the invention can be equally applied to forward link transmissions but offers additional advantages when provided in the remotemobile station. In addition, it will be understood by one skilled in the art that the control data can be multiplexed onto other channels transmitted by the remote station. However, in the preferred embodiment, the control data is multiplexed onto the pilot channel because unlike the fundamental and supplemental channels, the pilot channel is always present regardless of whether the remote station has traffic data to send to the central communications station. In addition, although the present invention is described in terms of multiplexing the data onto the pilot channel, it is equally applicable to the case where the power control data is punctured into the pilot channel.

Pilot data which consists solely of a stream of binary “1” values are provided to multiplexer (MUX) 300. In addition the control channel data, which in the exemplary embodiment is power control data consisting of +1 and −1 values indicative of instruction for the base station to increase or decrease its transmission power, are provided to MUX 300. Multiplexer 300 combines the two data streams by providing the control data into predetermined positions in the pilot data. The multiplexed data is then provided to a first input of multipliers 310 and 328.

The second input of multiplier 310 is provided with a pseudonoise (PN) sequence of +1 and −1 values. The pseudonoise sequence provided to multipliers 310 and 312 is generated by multiplying the short PN sequence (PN_(I)) by the long code. The generation of short PN sequences and long code sequences is well known in the art and described in detail in the IS-95 standard. The second input of multiplier 328 is provided with a pseudonoise (PN) sequence of +1 and −1 values. The pseudo noise sequence provided to multipliers 318 and 328 is generated by multiplying the short PN sequence (PN_(Q)) by the long code.

The output of multiplier 310 is provided to a first input of multiplier 314. The output of multiplier 318 is provided to delay element 320 which delays the input data by a time interval equal to half a chip. Delay element 320 provides the delayed signal to the subtracting input of subtractor 314. The output of subtractor 314 is provided for transmission to baseband filters and pilot gain elements (not shown).

The output of multiplier 328 is provided to delay element 330 which delays the input data by half a chip cycle as described with respect to delay 320. The output of delay element 330 is provided to a second summing input of summer 322. The first input of summing element 322 is the output of multiplier 312. The summed output from summer 322 is provided for transmission to baseband filters and pilot gain elements (not shown).

Traffic data to be transmitted on the supplemental channel, consisting of +1 and −1 values, is provided to a first input of multiplier 302. The second input of multiplier 302 is provided with a repeating Walsh sequence (+1, −1). As described above the Walsh covering is to reduce the interference between channels of data transmitted from the remote station. The product data sequence from multiplier 302 is provided to gain element 304 which scales the amplitude to a value determined relative to the pilot/control channel amplification. The output of gain element 304 is provided to a first input of summer 316. The output of summer 316 is provided to the inputs of multipliers 312 and 318 and processing continues as described above.

Traffic data to be transmitted on the fundamental channel, consisting of +1 and −1 values, is provided to a first input of multiplier 306. The second input of multiplier 306 is provided with a repeating Walsh sequence (+1,+1,−1,−1). As described above the Walsh covering reduces the interference between channels of data transmitted from the remote station. The product data sequence from multiplier 306 is provided to gain element 308 which scales the amplitude to a value determined relative to the pilot/control channel amplification. The output of gain element 308 is provided to a second input of summer 316. The output of summer 316 is provided to the inputs of multipliers 312 and 318 and processing continues as described above.

Referring to FIG. 9, the present invention is illustrated to include the necessary filtering operations and illustrates an additional benefit attained by combining the pilot and control data. That is a reduction in the amount of necessary filtering circuitry. As described with respect to FIG. 8, the pilot data and control channel data are multiplexed together by multiplexer (MUX) 350. The multiplexed data, consisting of +1 and −1 values, is provided to a first input of multipliers 352 and 354. The second input of multiplier 352 is provided by multiplying the short PN code PN_(I) by the long code in multiplier 390. The product from multiplier 352 is provided to finite impulse response (FIR) filter 356. In the exemplary embodiment, FIR 356 is a 48 tap FIR filter, the design of which is well known in the art. The second input of multiplier 354 is provided by multiplying the short PN code PN_(Q) by the long code in multiplier 392. The output of FIR 356 is provided to the summing input of subtractor 374. The output of subtractor 374 is provided for transmission to upconverters and pilot gain elements (not shown).

The product from multiplier 354 is provided to finite impulse response (FIR) filter 358. In the exemplary embodiment, FIR 358 is a 48 tap FIR filter, the design of which is well known in the art. It should be noted that by combining the pilot and power control data, two FIR filters have been eliminated since each channel requires two FIR filters. Elimination of two FIR filters reduces complexity, power consumption and chip area. The output of FIR 358 is provided to delay element 360 which delays the output by half a chip before providing the signal to a first summing input of summer 376. The output of summer 376 is provided for transmission to upconverters and pilot gain elements (not shown).

The supplemental channel traffic data consisting of +1 and −1 values are provided to a first input of multiplier 362. The second input to multiplier 362 is a repeating Walsh sequence (+1,−1) which as described previously reduce interference between the channels. The output of multiplier 362 is provided to a first input of multipliers 364 and 366. The second input of multiplier 364 is the pseudonoise sequence provided from multiplier 392 and the second input to multiplier 366 is the pseudonoise sequence provided from multiplier 390.

The output from multiplier 364 is provided to FIR/gain element 368 which filters the signal and amplifies the signal in accordance with a gain factor relative to unity gain of the pilot/control channel. The output of FIR/gain element 368 is provided to delay element 372. Delay element 372 delays the signal by ½ a chip before providing the signal to a first subtracting input of subtracting element 374. Processing of the output of subtractor 374 proceeds as described above.

The output from multiplier 366 is provided to FIR/gain element 370 which filters the signal and amplifies the signal in accordance with a gain factor relative to unity gain of the pilot/control channel. The output of FIR/gain element 370 is provided to a second input of summing element 376. Processing of the output of subtractor 376 proceeds as described above.

The fundamental channel traffic data consisting of +1 and −1 values is provided to a first input of multiplier 388. The second input to multiplier 388 is a repeating Walsh sequence (+1,+1,−1,−1) which as described previously reduces interference between the channels. The output of multiplier 388 is provided to a first input of multipliers 378 and 384. The second input of multiplier 378 is the pseudonoise sequence provided from multiplier 392 and the second input to multiplier 384 is the pseudonoise sequence provided from multiplier 390.

The output from multiplier 378 is provided to FIR/gain element 380 which filters the signal and amplifies the signal in accordance with a gain factor relative to unity gain of the pilot/control channel. The output of FIR/gain element 380 is provided to delay element 382. Delay element 382 delays the signal by ½ a chip before providing the signal to a second subtracting input of subtracting element 374. Processing of the output of subtractor 374 proceeds as described above.

The output from multiplier 384 is provided to FIR/gain element 386 which filters the signal and amplifies the signal in accordance with a gain factor relative to unity gain of the pilot/control channel. The output of FIR/gain element 386 is provided to a third input of summing element 376. Processing of the output of subtractor 376 proceeds as described above.

Referring to FIG. 10, a receiver for processing the data wherein the control data is multiplexed with the pilot signal data is illustrated. The data is received by an antenna(not shown) and downconverted, filtered and sampled. The filtered data samples are provided to delay elements 400 and 402. Delay element 400 and 402 delay the data by half of a chip cycle before providing the data to a first input of multipliers 404 and 406. The second input of multipliers 404 and 406 are provided with a pseudonoise sequence provided by multiplier 450. Multiplier 450 generates the pseudonoise sequence by multiplying the short code PN_(I) by the long code as described previously.

The filtered samples are also provided directly (without delay) to a first input of multipliers 446 and 448. The second input of multipliers 446 and 448 are provided with a pseudonoise sequence by multiplier 452. Multiplier 452 generates the pseudonoise sequence by multiplying the short PN code (PN_(Q)) by the long code. The output from multiplier 404 is provided to a first input of summer 408, and the output from multiplier 446 is provided to a second input of summer 408. The output from multiplier 406 is provided to a summing input of subtractor 410, and the output from multiplier 448 is provided to a subtracting input of subtractor 410.

The output of summer 408 is provided to delay element 412 and pilot symbol selector 434. Pilot symbol selector 434 gates out the control data from the pilot data, before providing the signal to pilot filter 436. Pilot filter 436 filters the signal and provides the filtered pilot signal to multipliers 416 and 418. Similarly, pilot symbol selector 438 gates out the control data from the pilot data, before providing the signal to pilot filter 440. Pilot filter 440 filters the signal and provides the filtered pilot signal to multipliers 442 and 444.

Delay 412 is used to synchronize the data through the two paths, before they are provided to multiplier 416. That is to say that delay element 412 provides a delay that is equal to the processing delay of pilot symbol selector 434 and pilot filter 436 which is equal to the processing delay of pilot symbol selector 438 and pilot filter 440. Similarly delay element 414 synchronizes the data provided to multipliers 418 and 442.

The output of delay element 412 is provided to a first input of multipliers 416 and 444. The second input to multiplier 416 is provided by the output of pilot filter 436. The second input to multiplier 444 is provided by pilot filter 440. The output of delay element 414 is provided to a first input to multipliers 418 and 442. The second input to multiplier 418 is provided by the output of pilot filter 436. The second input to multiplier 442 is provided by pilot filter 440.

The output of multiplier 416 is provided to a first input of summer 420 and the second input to summer 420 is provided by the output of multiplier 442. The sum from summer 420 is provided to control symbol selector 424 which separates the control data from the pilot channel data and provides that information to a control processor not shown which adjusts the base station transmission power in response thereto.

The output from multiplier 418 is provided to a summing input of subtractor 422. The output from multiplier 444 is provided to a subtracting input of subtractor 422. The output of subtractor 422 is provided to a first input of multiplier 426. The second input of multiplier 426 is provided with the repeating Walsh sequence (+1,−1). The product from multiplier 426 is provided to summing element 428 which sums the input bits over the Walsh sequence period to provide the supplemental channel data. The output of subtractor 422 is provided to a first input of multiplier 430. The second input of multiplier 430 is provided with the repeating Walsh sequence (+1,+1,−1,−1). The product from multiplier 430 is provided to summing element 432 which sums the input bits over the Walsh sequence period to provide the fundamental channel data.

Thus, a multi-channel, high rate, CDMA wireless communication system has been described. The description is provided to enable any person skilled in the art to make or use the present invention. The various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without the use of the inventive faculty. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein. 

We claim:
 1. A method for generating modulated data for transmission from a subscriber unit to a base station, the method comprising: combining a pilot signal with a control signal to produce first channel data; multiplying a first data signal with a first code, said first code being orthogonal to said pilot signal, to produce second channel data; performing complex multiplication of said first channel data and said second channel data with a complex pseudonoise (PN) code, said complex PN code comprising an in-phase PN code and a quadrature-phase PN code.
 2. The method of claim 1 wherein said combining comprises multiplexing said pilot signal with said control signal.
 3. The method of claim 1 wherein said combining comprises puncturing said control signal into said pilot signal.
 4. The method of claim 1 wherein said complex multiplication is performed using said first channel data and said in-phase PN code as real components and using said second channel data and said quadrature-phase PN code as imaginary components.
 5. The method of claim 4 wherein said performing complex multiplication comprises delaying a product of said second channel data and said quadrature-phase PN code by a delay amount and delaying a product of said first channel data and said quadrature-phase PN code by said delay amount.
 6. The method of claim 5 wherein said delay amount is equal to half a chip.
 7. The method of claim 1 wherein said first code is a Walsh code.
 8. The method of claim 1 wherein said first code is the Walsh code +, −.
 9. The method of claim 1 wherein said first code is the Walsh code +, +, −, −.
 10. The method of claim 1 further comprising adjusting a gain of said second channel signal.
 11. A subscriber unit apparatus comprising: means for combining a pilot signal with a control signal to produce first channel data; means for multiplying a first data signal with a first code, said first code being orthogonal to said pilot signal, to produce second channel data; means for performing complex multiplication of said first channel data and said second channel data with a complex pseudonoise (PN) code, said complex PN code comprising an in-phase PN code and a quadrature-phase PN code.
 12. The apparatus of claim 11 wherein said means for combining comprises means for multiplexing said pilot signal with said control signal.
 13. The apparatus of claim 11 wherein said means for combining comprises means for puncturing said control signal into said pilot signal.
 14. The apparatus of claim 11 wherein said means for performing complex multiplication is configured to perform said complex multiplication using said first channel data and said in-phase PN code as real components and using said second channel data and said quadrature-phase PN code as imaginary components.
 15. The apparatus of claim 14 wherein said means for performing complex multiplication comprises means for delaying a product of said second channel data and said quadrature-phase PN code by a delay amount and delaying a product of said first channel data and said quadrature-phase PN code by said delay amount.
 16. The apparatus of claim 15 wherein said delay amount is equal to half a chip.
 17. The apparatus of claim 11 wherein said first code is a Walsh code.
 18. The apparatus of claim 11 wherein said first code is the Walsh code +, −.
 19. The apparatus of claim 11 wherein said first code is the Walsh code +, +, −, −.
 20. The apparatus of claim 11 further comprising gain adjustment means for adjusting a gain of said second channel signal.
 21. A subscriber unit apparatus comprising: multiplexer for multiplexing a pilot signal with a control signal to produce first channel data; mixer for multiplying a first data signal with a first code, said first code being orthogonal to said pilot signal, to produce second channel data; complex spreader for performing complex multiplication of said first channel data and said second channel data with a complex pseudonoise (PN) code, said complex PN code comprising an in-phase PN code and a quadrature-phase PN code.
 22. The apparatus of claim 21 wherein said complex spreader is configured to perform said complex multiplication using said first channel data and said in-phase PN code as real components and using said second channel data and said quadrature-phase PN code as imaginary components.
 23. The apparatus of claim 22 wherein said complex spreader comprises means for delaying a product of said second channel data and said quadrature-phase PN code by a delay amount and delaying a product of said first channel data and said quadrature-phase PN code by said delay amount.
 24. The apparatus of claim 23 wherein said delay amount is equal to half a chip.
 25. The apparatus of claim 21 wherein said mixer is configured to utilize a Walsh code as said first code.
 26. The apparatus of claim 21 wherein said mixer is configured to utilize the Walsh code +, − as said first code.
 27. The apparatus of claim 21 wherein said mixer is configured to utilize the Walsh code +, +, −, − as said first code.
 28. The apparatus of claim 21 further comprising a gain element for adjusting a gain of said second channel signal. 